Circuit Labs - Electronic Circuits Collection

This ultra-simple circuit will produce the familiar sound of sirens used by US police cars on emergency calls. A small lamp will also flash synchronously with the siren sound. The circuit is capable of powering loads greater than 1 A for one or more lamps or a powerful loudspeaker, the kit producing quite a bit of noise and light.

The circuit is built from two astable multivibrators, in this case the familiar 555 of which two are present in an NE556 case. Of course, you are free to use two 555s if that suits you better. Both timer ICs are configured to operate as astable multivibrators.

The first timer is configured with R1, R2 and C2 to supply a rectangular signal of about 2 Hz at pin 9. The lamp is switched on and off by way of power transistor T1. The second 555 is configured using R4, R5 and C5, and supplies a square wave at pin 5 that drives the loudspeaker. The toggling voltage at the output of the first timer (pin 9) causes electrolytic capacitor C3 to be partly charged and discharged, periodically, via resistor R3. C3 is connected to the control input of
the second timer (pin 3), causing it to work as a VCO (voltage controlled amplifier). The upshot is that the frequency of the square wave applied to the loudspeaker rises and falls periodically, rendering a good imitation of the wailing sound of the US police car siren (we hear too often in movies).

The small number of dead-standard components used enables this circuit to be built on Veroboard without problems.

Author: Arthur Schilp

(Elektor Electronics Magazine – 2006)

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Energy is becoming more and more expensive, and so we are always on the lookout for ways to save energy in circuits. The author has decided to look at how to recovery energy from a relay switching circuit. If a relay is driven by a transistor switching stage it is usual to connect a flywheel diode in parallel with the coil to short out the back EMF produced when the relay current is switched off (Figure 1).

If an LED is wired in series with the flywheel diode (Figure 2) it will flash every time there is an inductive spike with the transistor turns off. The duration and brightness of the flash (and indeed, whether the spike is large enough to destroy the LED!) depend on the rate of change of the current in the relay coil and its inductance:

ui = –L di / dt

So far we have not actually recovered any energy. Figure 3 shows a theoretical design where the energy stored in the relay coil is recovered so that it can be used to supply a (low-power) circuit.

The greater the inductance of the coil and the more frequently it is switched, the more energy is stored in capacitor C. The zener diode (in series with the flywheel diode) limits the maximum voltage to which the capacitor can be charged. Measured relative to ground the open-circuit voltage at point A is the sum of the capacitor voltage due to the recovered energy and the supply voltage. In particular, the voltage at point A is higher than the supply voltage.

The author would be interested in discussing these ideas further with readers. His e-mail address is info@peterlay.de.

Author: Peter Lay

(Elektor Electronics Magazine – 2006)

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Integrated opamps offer advantages such as ease of use, good price-performance ratio and small physical size. However, they seldom have an output current drive capability of greater than about 12 mA, and so they are not suitable for use in 20 mA current loop applications, for example. One solution is to add a driver stage with the necessary output power, comprising perhaps two to four transistors and a number of other components.

This design takes up board space and is relatively expensive, tending to offset the advantages of the integrated device. An alternative possibility is to boost the output drive capability by connecting opamps in parallel. The output current will then be approximately proportional to the number of opamps. Instead of a single opamp a dual or quad device is used to achieve greater output power.

The idea is shown in Figure 1. The output of the first opamp is connected to the input of a further noninverting opamp stage as well as being connected to the output of the circuit via a resistor. The first opamp thus drives the second non-inverting amplifier which provides all the output current of the circuit as long as that remains within its normal capability.

As the output current demand increases the second opamp will reach the limit of its drive capability. Its gain will then fall off and a voltage difference will develop across its inputs. The first opamp will then start to deliver more and more current to the output via the resistor, and the sum of the output currents of the two opamps thus flows through load resistor RL.

By adding another resistor we can compare the current contributions from the two opamps (Figure 2). The complete circuit with two opamps is shown in Figure 3. The principle of the circuit can be extended to more opamps with their output currents being added together (see Figure 4).

Author: Klemens Viernickel

(Elektor Electronics Magazine – 2006)

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When dealing with relays we distinguish between the pull-in voltage and the hold voltage. Depending on the type of device, the latter is from about 10 % to 50 % lower than the former. This means that once we have safely pulled in the relay armature we can drop the coil voltage to its hold value, thus reducing the power dissipated. The simple circuit shown here does just that: it consists of a parallel combination of an LED, an electrolytic capacitor and a resistor, together placed in series with the relay coil. As well as saving energy by reducing the operating current of the relay and increasing its operating life the circuit also has the advantage of providing a status indicator in the form of the LED.

The author has used this circuit with practically all types of relay, with various rated currents and voltages. The recommended component values are as follows:

• The electrolytic capacitor should have a value between 100 μF and 1000 μF with a working voltage of 6.3 V, depending on the rated current of the relay coil.

• The resistor value should be between 10 Ω and 1 kΩ so that in the active state a current of 20 mA flows through the LED.

• A standard green or yellow LED with a rated forward current of 20 mA should be used. When using relays with a very low coil current low-current LEDs may be substituted. Add a zener diode in series for higher coil voltages such as 24 V or 48 V.

Author: Klemens Viernickel

(Elektor Electronics Magazine – 2006)

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This circuit demonstrates how easily a timebase can be designed using a minimum number of components. This circuit consists of no more than an IC and an oscillator to which a few connections and decoupling parts are added. Use is made of a single IC from the 4000 series, the type 4521 oscillator / counter with 24 steps. Here we only use the counter section of the IC.

The IC is supplied with a clock signal by an oscillator module. The clock signal is divided down by the 4521 to obtain certain values. Discrete frequencies obtained in this way are available on the counter output pins of the 4521. For example, pin 10 (Q18) supplies the clock frequency (applied to pin 6) by a factor of 218 or 262,144. Likewise pin 1 (Q24) divides the input signal by 224 or 16,777,216.

By using a clock frequency of Accurate timebase 2.097152 MHz the following timebase frequencies are obtained at the output of the 4521:

• Pin 10 (Q18): 8 Hz;
• Pin 11 (Q19): 4 Hz;
• Pin 12 (Q20): 2 Hz;
• Pin 13 (Q21): 1 Hz;
• Pin 14 (Q22): 0.5 Hz;
• Pin 15 (Q23): 0.25 Hz;
• Pin 1 (Q24): 0.125 Hz.

In case other frequencies are required, a different crystal oscillator module should be selected. For flexibility, an IC socket is recommended so modules can be exchanged quickly. Alternatively, modules may be connected to the 4521 by means of a selector switch.

To keep the circuit as versatile as possible, the outputs of the 4521 counter may be made available for connection to the outside world by way of a pinheader. For the sake of convenience, the pinheader also supplies +5 V and GND.

Author: Thomas Pototschnig

(Elektor Electronics Magazine – 2006)

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The humble PCB pin is used not only to connect wires to PCB tracks, but also for test and measurements points on circuit boards. Despite their mechanical simplicity, PCB pins are surprisingly difficult to mount and solder. Frequently encountered problems (admit it!) are burnt fingers while soldering the pin, and pins dropping from the board when the board is turned, or, worse, when they’re being soldered. Try to keep it in place with your finger — too hot to touch — look for pliers — pin dislocated — and so on.

It’s a good idea to drill the holes for the PCB pins a fraction smaller than the pin diameter — that way, the pin remains in place when you’re ready to solder it. However, that also requires a bit more force to push the pin into position and a tool is then useful to prevent injuring your fingers. Pliers may be less suitable, especially if considerable force is required on the pin. To the best of our knowledge, no commercial tool exists for the purpose. No problem, let’s make a PCB pin insertion tool ourselves.

A discarded round file or screwdriver is great. Cut or grind the shaft as straight as you can and drill a hole in the end so the ridge on the PCB pin is secure against the surface of the tool. The PCB pin is inserted into the tool and held in place with a nail or fingertip. The tool handle provides a secure grip and allows considerable insertion force to be applied. If an old file is used, it is recommended to cover the remaining section of the steel surface with heat shrink tubing. This will prevent injury to your hand.

Author:  Luc Lemmens (Elektor labs)

(Elektor Electronics Magazine – 2006)

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Even if a large number of album titles once available on vinyl are now, little by little, being proposed as CDs, not all are available and far from it. You may have treasures in your collection that you would like to burn on CDs. First, preserving a CD is easier than preserving a vinyl record, and second, we have to admit that turntables are disappearing, even on fully-equipped Hi-Fi systems. From a point of view of software and PCs, converting from vinyl to CD is not a problem. A large number of programs, whether paid for freeware, are available to re-master vinyl records with varying degrees of success and to eliminate pops, crackles and other undesirable noises. All of these programs work with the sound card of your PC and that, admittedly, is where the problem starts. Most high-quality turntables are equipped with a magnetic cartridge which typically delivers just a few mV. The cartridge signal requires a correction of a specific frequency, called RIAA correction.

If our older readers will perfectly recall what RIAA is all about, others from the CD generation may not know what the acronym RIAA stands for, guessing it may have something to do with illegal downloading of music on the Internet. For mechanical reasons related to the vinyl engraving procedure, high-boost frequency correction is carried out while respecting a very precise curve defined a long time ago by the RIAA (Recording Industry Association of America) and, which therefore, quite naturally, was baptized RIAA correction.

Reversing the correction is the role of to the preamplifier for the magnetic cartridge. Since this correction boosts the lowest frequencies, such a preamplifier is very sensitive to all undesirable noises, hums, including, of course, the one coming from the 50-Hz (or 60-Hz) mains power supply. It is important to take that into account while making this project which must be done carefully with respect to grounding and shielding. The schematic of our preamplifier is very simple because it uses a very low-noise dual operational amplifier. Here the NE5532 is used, whose response curve is modelled by R7, R8, C8, and C9 (or R14, R15, C13, and C14 respectively) in order to match the RIAA correction as closely as possible.

The input has an impedance of 47 kΩ, which is the standardized value of magnetic cartridges, and its 1,000-Hz gain is 35 dB which allows it to supply an output level of a few hundred mV typically required by for the line input of a PC sound cards.

The connection between the cartridge and the input of the amplifier requires shielded wiring to avoid the hum problems discussed above. Likewise we recommend fitting the assembly in a metal housing connected to the electric ground. With respect to the power supply, three solutions are proposed: If you are a purist and you want to rule out any noise whatsoever, you will utilize a simple 9-V battery. Then, the components outlined with a dotted line will not be useful. Since the circuit only uses a few mA, such a solution is acceptable unless your collection of vinyls is impressive...

If you desire a more elegant technical solution that might sometimes cause more undesirable noise on the signals, you may want to wire up the components within in the dotted lines and you can steal the 12 V positive voltage available from your PC. A Y-connector inserted on the power supply of one of the internal drives or peripherals will work very well for that.

Finally, you may also use a mains adapter set to 12 V and connect it to the +12-volt point of the drawing in order to benefit from additional filtering, which is not a luxury for some.

Author: Christian Tavernier

(Elektor Electronics Magazine – 2006)

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There are amplifier circuits that have capacitance between the input and output. If the gain is positive, this can lead to oscillations. If the gain is negative, another outcome is the result. We can deduce this from the following theoretical circuit.

An amplifier with a negligibly low output impedance, an infinitely high input impedance and gain A has feedback in the form of capacitor C (refer Figure 1). The gain A is negative. In addition, input current I, input voltage U and output voltage Uo are also drawn in. The input current I is equal to Ic and the input voltage U is equal to Uc + Uo. Uo in turn is equal to the product A U. From this follows that

Uc = U–Uo = U (1–A).

Substituting into the formula the current that flows through a capacitor, Ic = C(dUc/dt) results in

I = C.dU(1-A)/dt

We rearrange this as  I=(1-A).C.dU/dt

Now we can see that the gain determines the relationship between I and C. C appears to be larger by a factor of (1–A) (note: if A is negative, you can actually speak of a factor 1+A larger).

This is called the Miller effect. The apparent (larger) capacitance is called the Miller capacitance. When designing signal amplifiers you need to take this capacitance into account. We can actually use this Miller capacitance in other ways. If we make A variable, with an adjustable resistor for example, we create a variable capacitor. For this purpose we conceived the following schematic (see Figure 2).

Cm is the apparent capacitor between the input of the circuit and ground. If we connect a signal generator via a series resistor to the input and measure the input voltage with an oscilloscope, we can easily determine the corner frequency.

JFET-opamp A1 is necessary to prevent R1 from appearing in parallel with Cm and affecting the corner frequency. A2 is the actual (inverting) amplifier. The gain of A2 is equal to P1/R1. C is the capacitor which is enlarged artificially. The remaining components only serve to set the operating point of the circuit.

Cb blocks any DC voltages and needs to be relatively large, for example 25 times the maximum Cm. From the test results it appears that Cm is indeed equal to (1+P/R1) C. Cm can be varied with a potentiometer from about 560 pF to 12 nF.

As is usually the case, there are a few limitations in practice. The input signal may not be too large. Otherwise, the AC voltage across Cm will cause clipping at the output of the second opamp. At maximum Cm, the gain of A2 is about 20 times. The peak-to-peak value of the input voltage may therefore not be more than about 1/20 of the power supply voltage. The circuit will always work well for smaller signals, provided the frequency is not too high.

For A1 and A2 we used an LF356 and TL081 respectively. These are mainly used for frequencies not exceeding 100 kHz. Very fast JFET opamps could extend the useful frequency range to applications in the RF-range. For LF applications we could also use a dual
opamp for A1 and A2, such as the TL082.

The value of capacitor C can be changed to suit the application. With opamps of the type AD8099 with a C of 22 pF we can make a (tuning) capacitor with a value from 22 to 440 pF, for use up to 30 MHz. The alternative, a varicap diode that can be varied in capacitance over a range of 20 times (or more) is not used in practice very much any more. Other applications for this circuit are, for example, adjustable LC-filters for audio applications.

Author: Gert Baars

(Elektor Electronics Magazine – 2006)

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Although it does not have the same charm as real mercury barometers with long glass tubes on pieces of carved and polished wood, the Torricelli barometer discussed here is a functional equivalent and electronic replica of the Torricelli barometer.

Actually, rather than displaying the atmospheric pressure on the traditional digital displays, we preferred to reproduce the general look of this respected predecessor of electronic barometers. The mercury tube is, of course, replaced by a simple LED scale which, if not as beautiful, is still less toxic for the environment in case of breakage. As indicated on the drawing, the pressuresensor utilized is a Motorola MPX2200AP.

This circuit is adapted for measuring absolute pressure and has a range well suited for atmospheric pressure. Without entering too deep into the technical details, such sensors deliver an output of voltage proportional not only to the measured pressure but, unfortunately, to their supply voltage as well. Hence they must be powered from a stable voltage which is ensured here by the use of IC1.

Since the output of the MPX2200 is differential and at a very low level, we had to resort to the use of four operational amplifiers IC4.A to IC4.D, contained in one LM324, to obtain levels that can be processed easily. As long as potentiometer P1 is adjusted correctly, this group of operational amplifiers delivers a voltage of 1 volt per atmospheric pressure of 1,000 hPa to the LM3914. Since the atmospheric pressure will be within the range 950 to 1040 hPa at sea level, we need to make an expanded-scale voltmeter with this LM3914 in order to better exploit the 10 LEDs that it can control. That is the role of resistors R7 and R8 which artificially raise the minimum voltage value the chip is capable of measuring.

Consequently, we can ‘calibrate’ our LED scale with one LED per 10 hPa and thus benefit from a measurement range which extends from 950 hPa to 1040 hPa. In principle, you should not have a need to go beyond that in either direction. The circuit may be conveniently powered
from a 9-volt battery but only if used very occasionally. Since this is usually not the case for a barometer, we advise you to use a mains adaptor instead supplying approximately 9 volts.

Calibration basically entails adjusting the potentiometer P1 to light the LED corresponding to the atmospheric pressure of your location at the time. Compare with an existing barometer or, even better, telephone the closest weather station. They will be happy to give you the information.

* After Evangelista Torricelli, 1608-1647, Italian physician who proved the existence of atmospheric pressure and invented the mercury barometer.

Author: Christian Tavernier

(Elektor Electronics Magazine – 2006)

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It’s good fun to keep an eye on all sorts of things in your environment and on the basis of events in this environment to switch, for example, lamps or buzzers. To help with this, the light barrier described here can be used to guard an entrance. You can use it to signal of
someone is walking through the corridor, or to check if the car has been parked far enough in the garage to be able to close the door. The circuit consists of a transmitter, which sends modulated infrared light and a receiver, which recognizes this. The circuit used here is almost insensitive to daylight or fluorescent light and therefore can be used outside.

The transmitter (Figure 1) generates about 1000 times per second, for a period of 540 ms, a burst of 36 kHz. IC1 has been set with C1, R1 and R2 to a frequency of about 1000 Hz. The output of IC1 ensures that IC2 will oscillate about 1000 times per second for a period of about 540 ms. IC2 is set to a frequency of 36 kHz with C2, P1, R4 and R5. The output of IC2 drives the IR LED D1 via transistor T1. C3 and R3 prevent the reasonably high current through D1 from generating too much interference on the power supply rail.

The receiver (Figure 2) is quite a simple design, because IC3 already does a lot of the work for us. When the IC ‘sees’ an IR signal with a frequency of 36 kHz, the output of IC3 will become ‘0’. The transmitter circuit alternates between sending an IR-signal of 36 kHz for 540 ms and is quiet for 470 ms. When this signal arrives at IC3, C4 will discharge via D2. Because the non-inverting input of IC4a is set to 2.5 V, with the aid of R10 and R11, the output of IC4a will be a ‘1’. In the intervening quiet periods of 470 ms, C4 will partially charge via R8, but this is not of sufficient duration to exceed the voltage of 2.5 V. Only when the light barrier is interrupted will C4 charge far enough that the output of IC4a will toggle and become a ‘0’. Because IC4a has an open-collector output, C5 will be immediately discharged and the output of IC4b will become a ‘1’.

With R9 and C5 this signal is stretched to about one second. If you increase the value of R9 to 100 kΩ, then this will become about 10 seconds. R12 and R13 are included to prevent chatter of the output around the trigger point, although  there is not really a risk of that happening in this circuit. Together with R14, the output of IC4b delivers a clean logic signal that we can use for further processing. The quickest way of calibrating the frequency of IC2 to 36 kHz, using P1, is with the aid of an oscilloscope. If you do not have one of those, then point the IRLED D1 at the receiver IC3 and turn P1 so that the voltage on the inverting input of IC4a is as low as possible. Make sure that IC3 during the calibration does not receive too high a signal by placing the IR-LED a considerable distance away or by not pointing directly at the receiver. If this procedure is not that successful then just set P1 to the centre position, this works just fine usually.

You should not have a problem with ambient light with this circuit. If you do have a problem because, for example, there is direct sunlight on IC3, then you will need to place it inside a small tube and point it at the IR LED. In this way no direct sunlight can reach the receiver. If
the IR LED and the receiver are placed too close together it is possible that the receiver will sense light reflected off the walls, even when someone is standing between the transmitter and receiver. In this case the solution is also a short piece of tube for both the transmit LED as well as the receiver (Figure 3). Make sure that the tubes are opaque (paint black or use water pipe, for example). The wires to the IR LED can be several meters long without any problems. Do not place the receiver IC too far from the circuit.

Relate article: Direction Sensitive Light Barrier

Author: Heino Peters

(Elektor Electronics Magazine – 2006)

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With two light barriers closely positioned one after the other it is possible to establish in which direction they have been crossed. If, for example, you place it at the entrance of the toilet then you can use it to control the lights: on when entering and off when leaving the room. The circuit for this has many similarities with the modulated light barrier appearing elsewhere in this Summer Circuits issue.

There are two ways to position the light barriers, namely a completely duplicated installation in opposing directions (this to prevent mutual interference) and a version with one IR transmitter and two receivers. Both types of installation are shown here, which one is most suitable depends on the actual application.

When used in a doorway, one transmitter is sufficient if the receivers are placed about 5 cm apart. With a wider passage, an installation with two separate IR-transmitters is a better solution. This circuit has a range of several meters, even if the sun shines directly on the receiver! We use the exact same IR-transmitter(s) as for the modulated light barrier. For the installation with two separate IR-transmitters it is sufficient to duplicate R6, T1, D1, C3 and R7 from the circuit of the modulated light barrier. Output OUT (pin 3) of IC2 can drive two of these IR-drivers without any difficulty. The receivers are slightly different than those of the modulated light barrier and the circuit is the same for both types of installation.

We again use the TSOP1736, which is sensitive to IR-light that is modulated at a frequency of 36 kHz. D2, R8 and C4 ensure that the received pulses from IC3 at the output of IC5a result in a ‘1’ when the beam is not interrupted. When the beam is interrupted this output will become a ‘0’ within about 1 ms. In the same way IC5b generates a ‘0’ when IC4 stops receiving IR-light. The 4013 CMOS-IC used here contains two D-flipflops, of which we use only one. The instant that light barrier 2 (IC4) is unblocked again, is used to clock the state of light barrier 1 (IC3) through to output Q1. This signal drives the relay via T2, which operates the light in the
room. The circuit therefore turns the light on or off the moment that light barrier 1 is uninterrupted.

Relate article: Modulated Light Barrier

Author: Heino Peters

(Elektor Electronics Magazine – 2006)

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The losses in a bridge rectifier can easily become significant when low voltages are being rectified. The voltage drop across the bridge is a good 1.5 V, which is a hefty 25% with an input voltage of 6 V. The loss can be reduced by around 50% by using Schottky diodes, but it
would naturally be even nicer to reduce it to practically zero. That’s possible with a synchronous rectifier. What that means is using an active switching system instead of a ‘passive’ bridge rectifier.

The principle is simple: whenever the instantaneous value of the input AC voltage is greater than the rectified output voltage, a MOSFET is switched on to allow current to flow from the input to the output. As we want to have a full-wave rectifier, we need four FETs instead of four diodes, just as in a bridge rectifier.

R1–R4 form a voltage divider for the rectified voltage, and R5–R8 do the same for the AC input voltage. As soon as the input voltage is a bit higher than the rectified voltage, IC1d switches on MOSFET T3. Just as in a normal bridge rectifier, the MOSFET diagonally opposite T3 must also be switched on at the same time. That’s taken care of by IC1b. The polarity of the AC voltage is reversed during the next half-wave, so IC1c and IC1a switch on T4 and T1, respectively.

As you can see, the voltage dividers are not fully symmetrical. The input voltage is reduced slightly to cause a slight delay in switching on the FETs. That is better than switching them on too soon, which would increase the losses. Be sure to use 1% resistors for the dividers, or (if you can get them) even 0.1% resistors.

The control circuit around the TL084 is powered from the rectified voltage, so an auxiliary supply is not necessary. Naturally, that raises the question of how that can work. At the beginning, there won’t be any voltage, so the rectifier won’t work and there never will be any voltage... Fortunately, we have a bit of luck here. Due to their internal structures, all FETs have internal diodes, which are shown in dashed outline here for clarity. They allow the circuit to start up (with losses). There’s not much that has to be said about the choice of FETs – it’s not critical. You can use whatever you can put your hands on, but bear in mind that the loss depends on the internal resistance.

Nowadays, a value of 20 to 50 mW is quite common. Such FETs can handle currents on the order of 50 A. That sounds like a lot, but an average current of 5 A can easily result in peak currents of 50 A in the FETs. The IRFZ48N (55 V @ 64 A, 16 mW) specified by the author is no longer made, but you might still be able to buy it, or you can use a different type. For instance, the IRF4905 can handle 55 V @ 74 A and has an internal resistance of 20 mΩ.

At voltages above 6 V, it is recommended to increase the value of the 8.2-kΩ resistors, for example to 15 kΩ for 9 V or 22 kΩ for 12 V.

Author: Wolfgang Schubert

(Elektor Electronics Magazine – 2006)

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Light effects have always been popular. Now that LEDs are available in all sorts of shapes, sizes and colors for reasonable prices, a whole gamut of possibilities has become feasible. Examples are case modding (embellishing PC cases with all kinds of lights, windows, etc.), adorning scooters, motorcycles and cars with various light ornaments, mood lighting in different colors and we could go on.

In Elektor Electronics we also regularly feature circuits with LEDs. One circuit flashes LEDs, another drives multicolored LEDs. On one occasion standard logic (counters, shift registers, etc.) is used to drive the LEDs, on another occasion a microcontroller is used. But there are also solutions that do not require additional driving electronics.

Ordinary flashing LEDs that require no more than a series resistor have been around for donkey’s ages. They are quite nice, but spectacular they are certainly not. The company I.C. Engineering offers something much nicer: a three color LED in a package with a diameter of 5 mm, which also contains all the control electronics.

This ‘LED’ only requires a power supply voltage of 3 V to give a continuous ‘light show’. The colors blend slowly from one to another. This effect is even nicer if the components are used next to each other. Because of small variations between LEDs, one LED will change colour a little faster than another, which results in a colorful play of lights. This ‘LED’ is eminently suitable to make a nice light ornament without too much effort.

Author: _

(Elektor Electronics Magazine – 2006)

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The idea for this adapter was really born out of necessity. The 24.576-MHz crystal oscillator that is used in the Audio ADC 2000, (24 bit/96 kHz, March 2001) is not (easily) available any more. A colleague who was interested in the circuit and was keen to try out the prototype realised that a 25 MHz oscillator was used at the time. In order to create useful recording material it is of course necessary to use the correct sampling frequency, 48 kHz, that is. This requires 512 times 48000 Hz, or 24.576 MHz. Fortunately this frequency is available as part of a series of oscillators from Citizen, the CSX-750FC series, to be more specific.

These oscillators are housed in a very small SMD package. We originally used the SG531P-series from Seiko Epson in the design for the A/D-converter. This comes in a kind of 8-pin DIL package. So, to nevertheless enable us to use the Citizen version, we designed a very small circuit board that adapts the SMD device with 4 pins to the footprint for the 8-pin DIP version. The connection pin order is the same. In addition, we have made the PCB also suitable for the 14-pin version (SG531P series). This requires two additional pins. These are located at pins 7 and 8 of the 14-pin package and are connected to pins 4 and 5 respectively of the 8-pin package.

Pin 1 is in both cases the enable pin and pin 8 (8-pin) and 14 (14-pin) are +5 V. Pay close attention when ordering the oscillator. It so happens that there are also 3.3-V versions (CSX-750FB and FJ). You need a 5-V version for the Audio-DAC. There is also a third letter after the type number, which indicates the accuracy: C or F for 100 ppm and B for 50 ppm.

If the PCB is to be used in place of an 8-pin oscillator then you can trim the board along the line that is clearly visible on the solder side of the board. The solder side (copper side) is the top side. Just to be clear: the dot on the package of the CSX750FCC is pin 1 of the oscillator. We used thin pin headers for the connections so that the small adaptor can be fitted into an IC-socket or soldered directly onto a PCB. The IC is available from Digi-Key.

Author: Ton Giesberts

(Elektor Electronics Magazine – 2006)

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This circuit arose from the need of the author to provide a 5 V output from the 24 V battery of a solar powered generator. Although solar power is essentially free it is important not to be wasteful especially for small installations; if the battery runs flat at midnight you’ve got a long wait before the sun comes up again. The basic requirement was to make an efficient step-down converter to power low voltage equipment; the final design shown here accepts a wide input voltage from 9 to 60 V with an output current of 500 mA. The efficiency is very good even with a load of 1 mA the design is still better than a standard linear regulator.

The low quiescent current (200 μA) also plays a part in reducing losses. Some of the components specified (particularly the power MOSFET) are not the most economical on the market but they have been deliberately selected with efficiency in mind.

When power is applied to the circuit a reference voltage is produced on one side of R2. D1 connects this to the supply (pin 7) of IC1 to provide power at start-up. Once the circuit begins switching and the output voltage rises to 5 V, D2 becomes forward biased and powers
the IC from the output. Diode D1 becomes reverse biased reducing current through R1. When the circuit is first powered up the voltage on pin 2 of IC1 is below the reference voltage on pin 3, this produces a high level on output pin 6. The low power MOSFET T1 is switched on which in turn switches the power MOSFET T3 via R5 and the speed-up capacitor C4, the output voltage starts to rise.

When the output approaches 5 V the voltage fed back to the inverting input of IC1 becomes positive with respect to the non inverting input (reference) and switches the output of IC1 low. T1 and T3 now switch off and C3 transfers this negative going edge to the base of T2 which conducts and effectively shorts out the gate capacitance of T3 thereby improving its switch off time.

The switching frequency is not governed by a fixed clock signal but instead by the load current; with no load attached the circuit oscillates at about 40 Hz while at 500 mA it runs at approximately 5 kHz. The variable clock rate dictates that the output inductor L1 needs to have the relatively high value of 100 mH. The coil can be wound on ferrite core material with a high AL value to allow the smallest number of turns and produce the lowest possible resistance. Ready-made coils of this value often have a resistance greater than 1 Ω and these would only be suitable for an output load current of less than 100 mA.

The voltage divider ratio formed by R4 and R3 sets the output voltage and these values can be changed if a different output voltage is required. The output voltage must be a minimum of 1 V below the input voltage and the output has a minimum value of 4 V because of the supply to IC1.

A maximum efficiency of around 90 % was achieved with this circuit using an input voltage between 9 and 15 V and supplying a current greater than 5 mA, even with an input voltage of 30 V the circuit efficiency was around 80 %. If the circuit is used with a relatively low input voltage efficiency gains can be made by replacing D4 with a similar device with a lower reverse breakdown voltage rating, these devices tend to have a smaller forward voltage drop which reduces losses in the diode at high currents. At higher input voltage levels the value of resistor R1 can be increased proportionally to reduce the quiescent current even further.

Author: Michel Franke

(Elektor Electronics Magazine – 2006)

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Among the many anti-theft devices that are available, presence simulators have a special role to play. In fact, while an alarm system generally reacts the instant the intrusion is detected, or sometimes a little afterwards, in all cases the damage has already occurred. The purpose of the presence simulator is to stop intrusions beforehand by making crooks think that someone is at home. Working from the principle that the majority of home burglaries, with break-in, happen particularly at night, a properly designed presence simulator turns on the lights as evening falls, then turns them off a few hours later, causing an observer with bad intentions to believe that the premises are occupied.

Creating such a function with a microcontroller is certainly very easy and has already been done many times in the past, but the project we are proposing now is intended for those among you who do not want to, or who cannot program this type of circuit. As a result, our diagram only includes very common logic circuits from the CMOS 4000 family, with quite respectable results.

Ambient light is measured using the LDR R3 and, when it goes below a threshold determined by the adjustable potentiometer (P1) setting, like when night falls, it drives the IC1.A gate output to a low level. This has the effect of triggering triac T3 through gates IC1.C, IC1.D and transistors T1 and T2. At the same time, this clears the reset input from IC2 which is none other than the classic 4060 in CMOS technology.

Considering the values of C2, R4 and P2, the internal continuous oscillator in IC2 functions at a frequency on the order of 5 Hz. Consequently, its output Q12 (pin 2) changes state at the end of approximately one to two hours (depending on the P2 setting) while Q13 (pin 3) does the same, but in two to four hours. Depending on whether a link has been installed on S1 or on S2, gate IC1.B output thus changes state after one to four hours, having the effect of blocking triac TRI1 through IC1.D, T1 and T2. Simultaneously, diode D1 blocks the oscillator contained in IC2 and, therefore, the assembly stops in this state. It is dark, the light was lit for one to four hours, according to the setting of P2 and the wiring of S1 or S2, and it just went out. A return to the initial state can only happen after IC2 is reset to zero, which occurs when
its input from reset to zero (pin 12) goes to high level, in other words at dawn and LDR R3 detects lights again.

Thanks to its low consumption, this circuit can be directly powered by the mains using capacitor C4. The latter must be a class X or X2 model, rates for 230 VAC. Such a model, called a self-healing capacitor, is actually the only type of capacitor we should use for power supplies that are directly connected to the mains supply.

To ensure proper operation, we should pay careful attention to the placement of the LDR, to prevent the device being influenced not only by light from the house to be protected, but also by potential street lights, or even headlights of passing cars. Finally, since it is directly connected to the mains, the assembly must be mounted in an insulating housing, for obvious
safety reasons.

http://www.tavernier-c.com

Author: Christian Tavernier

(Elektor Electronics Magazine – 2006)

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Most homes today have at least a few infrared remote controls, whether they be for the television, the video recorder, the stereo, etc. Despite that fact, who among us has not cursed the light that remained lit after we just sat down in a comfortable chair to watch a good film? This project proposes to solve that problem thanks to its original approach. In fact, it is for a common on/off switch for infrared remote controls, but what differentiates it from the commercial products is the fact that it is capable of working with any remote control. Therefore, the first one you find allows you to turn off the light and enjoy your movie in the best possible conditions.

The infrared receiver part of our project is entrusted to an integrated receiver (Sony SBX 1620-52) which has the advantage of costing less than the components required to make the same function. After being inverted by T1, the pulses delivered by this receiver trigger IC2a, which is nothing other than a D flip-flop configured in monostable mode by feeding back its output Q on its reset input via R4 and C3. The pulse that is produced on the output Q of IC.2A makes IC.2B change state, which has the effect of turning on or turning off the LED contained in IC3. This circuit is an opto triac with zero-crossing detection which allows our setup to accomplish switching without noise. It actually triggers the triac T2 in the anode where the load to be controlled is found. The selected model allows us to switch up to 3 amperes but nothing should stop you from using a more powerful triac if this model turns out to be insufficient for your use.

In order to reduce its size and total cost, the circuit is powered directly from the mains using capacitor C5 which must be a class X or X2 model rated at 230 volts AC. This type of capacitor, called ‘selfhealing’, is the only type we should use today for power supplies that are connected to ground. ‘Traditional’ capacitors, rated at 400 volts, do not really have sufficient safety guarantees in this area. Considering the fact that the setup is connected directly to the mains, it must be mounted in a completely insulated housing. A power outlet model works very well and can easily be used to interspace between the grounded wall outlet
and that of the remote control device. Based on this principle, this setup reacts to any infrared signal and, as we said before, this makes it compatible with any remote control. On the other hand, it has a small disadvantage which is that sometimes it might react to the ‘normal’ utilization of one of these, which could be undesirable. To avoid that, we advise you to mask the infrared receiver window as much as possible so that it is necessary to point the remote control in its direction in order to activate it.

Author:  Christian Tavernier

(Elektor Electronics Magazine – 2006)

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This current-limiting circuit, shown in this example as part of a small bench power supply, could in principle be used in conjunction with any dual-rail current source. The part of the circuit to the left of the diagram limits the current at the input to the dual voltage regulator (IC4 to IC7) so that it is safely protected against overload. The circuit shown produces outputs at ±15 V and ±5 V.

The voltage regulators at the outputs (7815/7805 and 7915/7905) need no further comment; but the current-limiting circuit itself, built around an LM317 and an LM337, is not quite so self-explanatory. The upper LM317 (IC1) manages the current limiting function for the upper branch of the circuit. The clever part is the combination of the two resistors R1 and R3 between the output and the adjust input of the regulator. In the basic LM317 configuration in current-limiting mode (i.e., as a constant current source), just one resistor is used here, across which the regulator maintains a constant voltage of 1.25 V. The current is thus limited to a value of 1.25 V/R. To obtain a maximum current of 1 A, for example, the formula tells us that the necessary resistor value is 1.25 Ω. Unfortunately it is not practical to try to build an adjustable dual-rail current-limited supply in this way, as stereo potentiometers with a value of 1.2 Ω are extremely difficult, if not impossible, to obtain.

We can solve the problem using the technique of dividing the resistor into two resistors. Only the resistor at the output of the LM317 (R1) serves for current sensing. The second resistor (R3) causes an additional voltage drop depending on an additional (and adjustable) current. When the sum of the two voltages reaches 1.25 V current limiting cuts in. This makes it possible to adjust the current limit smoothly using the current in the second resistor (R3). This can be done simultaneously in the positive and negative branches of the circuit, as the diagram shows.

It would of course be wasteful to arrange for the current flowing in the second resistor to be of the same order of magnitude as the current in the main resistor. We therefore make the value of the second resistor considerably greater than that of the main one. If the main resistor (R1) has a value of 1.2 Ω (giving a maximum current of 1 A), and the second resistor (R3) a value of 120 Ω, the necessary voltage drop is achieved using an extra current of 10 ent limit will be 1 A. For the negative branch of the circuit the LM337, along with resistors R2 (1.2 Ω) and R5 (120 Ω), performs the same functions.

A further LM317 (IC3) is used to set the overall current limit point by controlling the additional current. The resistance used with this voltage regulator, wired as a current sink (R4 in series with P1) determines the additional current and therefore also the output current in both the negative and positive branches of the circuit. Since we also want the total resistance of R4 and P1 to be 120 Ω, we use a value of 22 Ω for R4 and 100 Ω for P1 to give a
wide adjustment range for the output current from a few milliamps to 1 A. The minimum input voltage for the circuit depends on the desired output voltage and maximum output current. The input to the 7815 should be at least 18 V. We should allow approximately a further 1.2 V + 2.2 V for the voltage drops across IC1 and R1. If we allow a total of 4 V for the current limiting circuit in each branch, this means that the circuit as a whole should be supplied with at least ±22 V to produce well-regulated outputs at ±15 V and ±5 V.

If the symmetrical input voltage is to be provided using a single transformer winding, two diodes and two smoothing capacitors, it important to ensure that the capacitor values are sufficiently large, as there will be considerably more ripple than there would be with full-wave rectification. Depending on the application, capacitors C6 to C9 at the outputs of the fixed voltage regulators can be electrolytics with a value of 4.7 μF or 10 μF. To improve stability, electrolytic capacitors can also be connected in parallel with C1, C2, C4 and C5.

Author: Malte Fischer

(Elektor Electronics Magazine – 2006)

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